Power convertor with low loss switching

ABSTRACT

A welding power supply includes an input rectifier that receives sinusoidal or alternating line voltage and provides a rectified voltage. A pre-regulator provides a dc bus and a convertor, such as a boost convertor, provides a welding output. The pre-regulator is an SVT (slow voltage transition) and an SCT (slow current transition) switched convertor. It may include a snubber circuit having a diode that is SVT switched. Also, the boost convertor may be SVT and SCT switched. The pre-regulator preferably includes a power factor correction circuit. The power source includes, in one embodiment, an inverter having a snubber circuit having a first switch in anti-parallel with a first diode, and a second switch in anti-parallel with a second diode. The first switch and first diode are connected in series with the second switch and the second diode, and the first and second switches are connected in opposing directions, to form a switched snubber.

This new application is a divisional of, and claims the benefit of thefiling date of, U.S. patent application Ser. No. 09/111,950, filed Jul.9, 1998, entitled Power Converter With Low Loss Switching now U.S. Pat.No. 6,115,273.

FIELD OF THE INVENTION

This invention relates generally to power sources used in welding and,more particularly, to welding power sources that have a pre-regulator.

BACKGROUND OF THE INVENTION

Power sources typically convert a power input to a necessary ordesirable power output tailored for a specific application. In weldingapplications, power sources typically receive a high voltage,alternating current, (VAC) signal and provide a high current weldingoutput signal. Around the world, utility power sources (sinuosidal linevoltages) may be 200/208 V, 230/240 V, 380/415 V, 460/480 V, 500 V and575 V. These sources may be either single-phase or three-phase andeither 50 or 60 Hz. Welding power sources receive such inputs andproduce an approximately 10-75 volt, DC or AC high current weldingoutput.

There are many types of welding power sources that provide powersuitable for welding, including inverter-based welding power sources. Asused herein, an inverter-type power supply includes at least one stagewhere DC power is inverted into ac power. There are several well knowninverter type power sources that are suitable for welding. These includeboost power sources, buck power sources, and boost-buck power sources.

Traditionally, welding power sources were designed for a specific powerinput. In other words, the power source cannot provide essentially thesame output over the various input voltages. More recently, weldingpower sources have been designed to receive any voltage over a range ofvoltages, without requiring relinking of the power supply. One prior artwelding power supply that can accept a range of input voltages isdescribed in U.S. Pat. No. 5,601,741, issued Feb. 11, 1997 to Thommes,and owned by the assignee of the present invention, and is herebyincorporated by reference.

Many prior art welding power supplies include several stages to processthe input power into welding power. Typically stages include an inputcircuit, a pre-regulator, an invertor and an output circuit thatincludes an inductor. The input circuit receives the line power,rectifies it, and transmits that power to the pre-regulator. Thepre-regulator produces a dc bus suitable for conversion. The dc bus isprovided to the invertor of one type or another, which provides thewelding output. The output inductor helps provide a stable arc.

The pre-regulator stage typically includes switches used to control thepower. The losses in switches can be significant in a welding powersupply, particularly when they are hard switched. The power loss in aswitch at any time is the voltage across the switch multiplied by thecurrent through the switch. Hard switching turn-on losses occur when aswitch turns on, with a resulting increase in current through theswitch, and it takes a finite time for the voltage across the switch todrop to zero. Soft switching attempts to avoid turn-on losses byproviding an auxiliary or snubber circuit with an inductor in serieswith the switch that limits the current until the transition to on hasbeen completed, and the voltage across the switch is zero. This isreferred to as zero-current transition (ZCT) switching.

Similarly, hard switching turn-off losses also occur when a switch turnsoff, with a resultant rise in voltage across the switch, and it takes afinite time for the current through the switch to drop to zero. Softswitching attempts to avoid turn-off losses by providing an auxiliary orsnubber circuit with a capacitor across the switch that limits thevoltage across the switch until the transition to off has beencompleted, and the current through the switch is zero. This is referredto as zero-voltage transition (ZVT) switching.

There are numerous attempts in the prior art to provide soft-switchingpower converters or invertors. However, these attempts often eithertransfer the losses to other switches (or diodes) and/or requireexpensive additional components such as auxiliary witches and theircontrol circuits. Thus, an effective and economical way of recovering(or avoiding) switching losses in power converters or inverters isdesirable. Examples of various attempts at soft switching are describedbelow.

U.S. Pat. No. 5,477,131, issued Dec. 19, 1995 to Gegner discloses a ZVTtype commutation. However, a controlled auxiliary switch and a coupledinductor are needed to implement the ZVT. Also, the primary current isdiscontinuous.

Some prior art designs require discontinuous conduction mode for dioderecovery. One such design is found in U.S. Pat. No. 5,414,613. This isundesirable because of excessive high frequency ripple in the powerlines.

Gegner also disclosed a ZVS converter that operated in a multi-resonantmode in U.S. Pat. No. 5,343,140. This design produced relatively highand undesirable RMS current and RMS voltage.

Another multi-resonant converter is disclosed in U.S. Pat. No.4,857,822, issued to Tabisz. This design causes undesirable high voltagestress during ZVS events and undesirable high current stress during ZCSevents.

U.S. Pat. No. 5,307,005 also requires an auxiliary switch. Losses occurwhen the auxiliary switch is turned off. This merely shifts switchinglosses, rather than eliminating them. Other designs that “shift” lossesare shown in U.S. Pat. Nos. 5,418,704 and 5,598,318.

A circuit that requires an auxiliary controlled switch but does not“shift” losses to the auxiliary switch is shown in U.S. Pat. No.5,313,382. This is an improvement over the prior art that shiftedlosses, but still requires an expensive controlled switch.

Another design that avoided “loss shifting” is shown in U.S. Pat. No.5,636,144. However, that design requires a voltage clamp for recoveryspikes, and 3 separate inductors. Also, the voltages on the inductors isnot well controlled.

A zero-current, resonant boost converter is disclosed in U.S. Pat. No.5,321,348. However, this design requires relatively complex magneticsand high RMS current in the switches and magnitudes. Also, a highreverse voltage is needed for the boost diodes.

When it is not practical or cost effective to use a true ZCT and ZVTcircuit, an approximation may be used. For example, slow voltage/currenttransitions (SVT and SCT) as used herein, describe transitions where thevoltage or current rise is slowed (rather than held to zero), while theswitch turns off or on.

A typical prior art welding power supply 100 with a pre-regulator 104and an output convertor or inverter 105 is shown in FIG. 1. An inputline voltage 101 is provided to a rectifier 102 (typically comprised ofa diode bridge and at least one capacitor). Pre-regulator 104 is ahard-switched boost converter which includes a switch 106 and aninductor 107. A diode 108 allows a capacitor 109 to charge up by currentflowing an inductor 107 when the switch 106 is turned off. The currentwaveform in inductor 107 is a rectified sinusoid with high frequencymodulation (ripple).

The amount of ripple may be reduced by increasing the frequency at whichswitch 106 is switched. However, as the frequency at which a prior arthard switched boost converter is switched is increased to reduce ripple,the switching losses can become intolerable.

Another drawback of some prior art power supplies is a poor powerfactor. Generally, a greater power factor allows a greater power outputfor a given current input. Also, it is generally necessary to have morepower output to weld with stick electrodes having greater diameters.Thus, a power factor correction circuit will allow a given welding powersupply to be used with greater diameter sticks for a given line power. Aprior art inverter that provided a good power factor is disclosed inU.S. Pat. Nos. 5,563,777. Many prior art convertors with power factorcorrection suffer from high switching losses. Examples of such prior artdesigns are found in U.S. Pat. Nos. 5,673,184; 5,615,101; and 5,654,880.

One type of known output convertor is a half-bridge, transformerisolated, inverter. However, such output invertors often have highswitching losses and/or require passive snubber circuits (whichincreases losses) because each snubber must operate in both directionsoverall, but only in one direction at a time. Also, known snubbercircuits generally have a limited range of acceptable loads and will notsnub proportional to the load, thus the losses are relatively high forlower loads.

Accordingly, a power circuit that provides little switching losses and ahigh (close to unity) power factor is desirable. Also, the pre-regulatorshould be able to receive a wide range of input voltages withoutrequiring relinking. A desirable output convertor will include a fullwave, transformer isolated, inverter, that is soft switch and has fullrange, full wave, low loss snubber.

SUMMARY OF THE PRESENT INVENTION

According to a first aspect of the invention a welding power supplyincludes an input rectifier that receives sinusoidal or alternating linevoltage and provides a rectified sinusoidal voltage. A pre-regulatorreceives the rectified input and provides a dc bus. An invertorconnector across the bus provides a welding output. The pre-regulator isan SVT (slow voltage transition) and an SCT (slow current transition)switched invertor.

In one embodiment the pre-regulator includes a snubber circuit havingdiode that is SVT switched.

In another embedment the inverter is a boost converter with a switch.The pre-regulator includes a snubber circuit having a capacitor and aninductor. The capacitor is connected to slow the switch voltage risewhile the switch is turning off, and the inductor is connected to slowthe switch current rise when the switch is turning on. The boostconverter includes a boost inductor, a switch, and an output capacitorin another embodiment. Also, the snubber includes a snubber capacitor, asnubber inductor, a first snubber diode, a second snubber diode, a thirdsnubber diode, a fourth snubber diode, and first and second snubbercapacitors. The snubber inductor, switch, an fourth diode are connectedsuch that current may flow from the boost inductor to any of the snubberinductor, switch, and fourth diode. Current flowing through the fourthdiode can flow through either the third diode or the second capacitor.Current flowing from the boost inductor through the snubber inductor canflow through either the first diode or the first capacitor. The fourthdiode and the second capacitor are connected across the switch andcurrent flowing through the third diode can flow through either thefirst capacitor and the snubber inductor or through the second diode.Current flowing through the fist and second diodes flows to the output.A fifth diode is connected in anti-parallel to the switch in oneembodiment.

A second aspect of the invention is a method of providing welding powerby rectifying a sinusoidal or alternating input line voltage andpre-regulating the sinuosidal input line voltage to provide a dc bus.The method further includes SVT and SCT switching a boost convertor. Thebus is converted to a welding output.

Pre-regulating includes, in one embodiment, maintaining a boostconverter switch off, and allowing current to flow through a boostinductor, a snubber inductor, and a first diode, to the dc bus, andturning the switch on and diverting current from the snubber inductor tothe switch. Current is reversed in the snubber inductor and a secondcapacitor is discharged through a first capacitor, a third diode, andthe snubber inductor, thereby transferring energy from the secondcapacitor to the snubber inductor. Current is diverted through a fourthdiode, the third diode and the first capacitor when the second capacitoris discharged, thereby transferring energy from the snubber inductor tothe first capacitor. The switch is turned off and current divertedthrough the fourth diode and into the second capacitor. Voltage on thesecond capacitor is allowed to rise until current begins to flow fromthe snubber inductor to the first capacitor and then current is divertedfrom the second capacitor through a third diode to the second diode. Thecurrent from the boost inductor to the snubber inductor increases untilall of the current from the boost inductor flows into the snubberinductor. Then current is diverted from the first capacitor to the firstdiode. This process is repeated.

One embodiment includes SVT turning off a diode in a snubber circuit.Another includes slowing the switch voltage with a capacitor rise whilethe switch is turning off, and slowing the switch current rise with aninductor while the switch is turning on to SVT and SCT switching a boostconvertor.

A third aspect of the invention is a welding power supply having aninput rectifier that provides a rectified voltage. A pre-regulatorreceives as an input the rectified signal and provides a dc bus. Aninvertor converts the bus to a welding output and the pre-regulatorincludes a power factor correction circuit.

Yet another aspect of the invention is a welding power supply having aninput rectifier and a preregulator, and an invertor. The pre-regulatorincludes a snubber circuit having a first switch in anti-parallel with afirst diode, and a second switch in anti-parallel with a second diode.The combination of the first switch and first diode are connected inseries with the combination of the second switch and the second diode,and the first and second switches are connected in opposing directions.

Another aspect of the invention is a welding power supply having aninverter with first and second current paths through a transformer, eachin a unique direction. The first current path includes at least a firstswitch with an anti-parallel first diode and the second current paththrough the transformer in a second direction, the second current pathincluding at least a second switch with an anti-parallel second diode. Asnubber includes a current path having a third switch with ananti-parallel third diode, a fourth switch with an anti-parallel fourthdiode. The third switch and anti-parallel diode are in series with, andoppositely directed from, the fourth switch and anti-parallel diode. Thesnubber also has a at least one snubber capacitor.

In alternative embodiments the first and second switches are in ahalf-bridge configuration or full bridge configuration. Also, thesnubber capacitor may be split into two capacitors.

Another aspect of the invention is a method of providing welding powerby turning on a first power switch and a first snubber switch, andallowing current to flow through the first power switch, an first dcbus, a first power capacitor, and in a first direction through atransformer. Then the first power switch is turned off and current flowsthrough the first snubber switch, a second snubber diode, a snubbercapacitor, and through the transformer in the first direction, while thefirst power switch is turning off, to provide a slow voltage transitionoff. Then current flows through a second anti-parallel power diode, asecond DC bus, a second power capacitor, and through the transformer inthe first direction, while the first power switch is continuing to turnoff, to continue providing a slow voltage transition off. The firstsnubber switch is also turned off. After the system is at rest a secondpower switch on and a second snubber switch are turned after the firstpower switch is off, and current flows through the second power switch,the transformer in a second direction, the second power capacitor, andthe second bus. The second power switch is turned off and current flowsthrough the second snubber switch, a first snubber diode, thetransformer in the second direction, and a snubber capacitor, while thesecond power switch is turning off, to provide a slow voltage transitionof. Then current flows through a first power diode, the transformer inthe second direction, and the first power capacitor, while the secondpower switch is turning off, to provide a slow voltage transition off.The second snubber switch is also turned off, and the process isrepeated.

Other principal features and advantages of the invention will becomeapparent to those skilled in the art upon review of the followingdrawings, the detailed description and the appended claims.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram of a prior art welding power supply having aboost convertor pre-regulator;

FIG. 2 is a block diagram of a welding power supply constructed inaccordance with the present invention;

FIG. 3 is a circuit diagram of a power factor correction circuit used inthe preferred embodiment;

FIG. 4 is a circuit diagram of the pre-regulator of FIG. 2;

FIGS. 5-13 are a circuit diagram of FIG. 4, showing various currentpaths;

FIG. 14 is a circuit diagram of a switching circuit;

FIG. 15 is a full wave inverter using the switching circuit of FIG. 14;

FIG. 16 is a control circuit; diagram; and

FIGS. 17-22 are the circuit diagrams of FIG. 15 showing various currentpath.

Before explaining at least one embodiment of the invention in detail itis to be understood that the invention is not limited in its applicationto the details of construction and the arrangement of the components setforth in the following description or illustrated in the drawings. Theinvention is capable of other embodiments or of being practiced orcarried out in various ways. Also, it is to be understood that thephraseology and terminology employed herein is for the purpose ofdescription and should not be regarded as limiting. Like referencenumerals are used to indicate like components.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

While the present invention will be illustrated with reference to awelding power supply using a boost converter for a pre-regulator andparticular circuitry, it should be understood at the outset that othercircuit topologies may be used, and the power supply may be used forother purposes, and still be within the intended scope of thisinvention.

A block diagram of a welding power supply constructed in accordance withthe preferred embodiment is shown in FIG. 2. Source 201 represents theinput line voltage used to provide power to the welding power supply.The input line voltage may be anywhere between 90 and 250 volts in thepreferred embodiment. The voltage typically operates at a frequency of60 hertz (in the United States) and is single phase in the preferredembodiment (although alternative embodiments used a three phase input).Other voltages may also be used.

The input voltage is provided to a rectifier 202, which may be a simplebridge rectifier. The output of rectifier 202 is a rectified sinusoid.

A pre-regulator 204 receives the rectified sinusoid from rectifier 202and provides a dc bus output to an output invertor 205. Pre-regulator204, in the preferred embodiment is a soft-switched boost convertorwhich provides close to a unity power factor. Other convertor andinvertor configurations may be used. Pre-regulator 204 also allows theinput voltage to be anywhere within a range of input voltages in thepreferred embodiment.

Convertor 205 is preferably a half-bridge, transformer isolated, soft(or slow) switched invertor. Such an output circuit will be described indetail below. Output convertor 205 may alternatively be a typicalforward convertor (generally a buck convertor and a transformer), andother output converters may be used in other embodiments. A circuitincluding an output buck convertor is described in U.S. patentapplication entitled Auxiliary Open Circuit Voltage Power Supply,invented by Vogel and Geissler, filed on even date herewith, (herebyincorporated by reference) and assigned to the assignee of thisinvention. The output of convertor 205 is provided through an inductor207 to welding output 208.

The circuit used in the preferred embodiment to implement pre-regulator204 is shown in FIG. 4 (along with rectifier 202 and voltage source201). The embodiment of FIG. 4 uses a 90-250 volt ac power line as inputvoltage 201. Rectifier 202 is comprised of diodes D6, D7, D8, and D9,which rectify the input voltage to provide a single polarity sinuosidalinput voltage.

The power factor correction portion (described below) of pre-regulator204 functions best when the input voltage is sinusoidal, although itcould be another alternating input. Thus, a small (10 μF) capacitor (notshown) is provided across input rectifier 202 in one embodiment tosmooth the input line voltage.

The rectified input voltage is applied to a boost inductor L1 (750 μH)which is connected with a boost switch Z1 (preferably an IGBT) to form aboost convertor. An anti-parallel diode D5 is connected across switch Z1to protect switch Z1 during transitions. The portion of the circuitwhich provides the lossless switching includes a snubber inductor L2(3.9 μH) a pair of capacitors C1 (1 μF) and C2 (0.068 μF), and diodesD1, D2, D3, and D4. Switch Z1 is switched in a known manner such thatthe output of pre-regulator 204 is a desired voltage, no matter what theinput voltage is. The output is provided across a capacitor C5 (2000 μF)that provides a stable voltage source (400 volts in the preferredembodiment) for the downstream convertor. Also, capacitor C5 preventsthe voltage from being dangerously high and damaging switch Z1.

The soft switching of pre-regulator 204 is best understood withreference to FIGS. 5-11, which show the circuit with various currentpaths (states). The first state (FIG. 5) is when switch Z1 is off, andthe current (arrow 501) is in a steady state condition through inductorsL1 and L2, and diode D1, to change output capacitor C5 (arrow 501).

Then, switch Z1 is turned on, and current from inductor L1 begins to bedirected through switch Z1 (arrow 501 of FIG. 6). Switch Z1 applies areverse voltage to inductor L2, causing its current to fall. Thus, thecurrent (in this state) is decreasing through inductor L2 and increasingthrough switch Z1. Inductor L2 effectively limits or slows the currentin switch Z1 at turn on until the switch voltage drops (to close tozero). Thus, the turn on has been a slow-current transition (SCT).

Eventually all of the current from inductor L1 flows through switch Z1,and current in inductor L2 drops until it becomes zero, and thenreverses. Capacitor C2 discharges through capacitor C1, diode D3, andinductor L2, as shown in FIG. 7, by arrow 701. Capacitors C1 and C2allow diode D1 to turn off with a SVT, thus reducing losses. Thedischarge occurs at a resonant frequency determined by the time constantof the inductance of inductor L2 and the series capacitance ofcapacitors C1 and C2 (f=1(2π(L2*(C1*C2/C1+C2)). The time it takes forcapacitor C2 to discharge is the SVT time for diode D1.

Capacitor C2 discharges to about zero volts, and diode D4 begins toconduct, as shown by arrow 801 in FIG. 8. When diode D4 conducts,inductor L2 releases the energy stored therein to capacitor C1 at aresonant frequency determined by inductor L2 and capacitor C1(f=1(2π(L2*C1))). The voltage energy on capacitor C1 is transferred tocurrent in inductor L2, and then to voltage on capacitor C1. The ratioof voltage transfer is nearly equal to the capacitance ratio.

When the charge transfer is complete, and current ceases to flow insnubber inductor L2, the snubber is reset, and current in inductor L1increases through switch Z1, as shown in FIG. 9. The circuit remains inthis state until the switch is turned off.

Next, switch Z1 is turned off, (FIG. 10) and current is diverted throughdiode D4 and into capacitor C2 (arrow 1001). Capacitor C2 provides theSVT time for switch Z1, thus a soft switching off is provided. Thevoltage on capacitor C2 continues to rise and eventually reaches the busvoltage (the voltage on capacitor C5) less the voltage on capacitor C1.

When this happens the voltage on capacitor C1 begins to reestablish thecurrent in inductor L2 (FIG. 11 and arrow 1101). The voltage oncapacitor C2 continues to rise until it reaches the bus voltage plus twodiode drops. At that time current from inductor L1 not taken by inductorL2 is diverted through diode D3 FIG. 12 and arrow 1201). The voltage oncapacitor C1 continues to increase the current in inductor L2.

Eventually all of the current from inductor L1 flows through inductorL2, and current through diodes D3 and D4 ceases (FIG. 13). Capacitor C1continues to give energy to the bus.

When all of the energy on capacitor C1 is expended (to the bus) currentflows from inductor L1 to inductor L2, and through diode D1. This is thestate initially described, with respect to FIG. 5, and the cyclerepeats.

Thus, the voltage rise across switch Z1 was slowed by capacitor C2 toallow the current to drop when switch Z1 was turned off. The currentrise in switch Z1 was slowed by inductor L2 to allow the voltage todrop, when switch Z1 was turned on. Moreover, diode D1 was soft-switchedby capacitors C1 and C2.

The portion of pre-regulator 204 that provides power factor correctionis a power factor correction circuit 404 (FIG. 4), and generally sensesthe input voltage waveform, and conforms the shape of the currentwaveform to be that of the line voltage waveform. This provides a powerfactor of very close to 1, 0.99 in the preferred embodiment. Powerfactor correction circuit 404 may be implemented using an integratedcircuit, such as a UC3854 or an ML4831, or with discrete components.Power factor correction circuit 404 receives as inputs the outputvoltage from rectifier 202, the output voltage from pre-regulator 204,and the output current of pre-regulator 204 (using a CT 405). Becausethe frequency of pre-regulator 204 (25 KHz) is much higher than that ofthe line (60 Hz) the pre-regulator current can be made to track theinput line voltage shape by sensing the shape of the input voltage, andcontrolling the input current in response thereto.

An embodiment of power factor correction circuit 404 having discreetcomponents is shown in FIG. 3 controls the switches so that the inputcurrent is shaped to match the input voltage, as well as regulates theDC bus.

The input voltage is rectified and provided to a 2 pole Bessel filterwhich removes switching frequencies. The Bessel filter includescapacitors 1602 (0.0022 μF) and 1603 (0.001 μF), resistors (1606-1608(1M ohms), resistors 1609-1610 (39.2K ohm), and an op amp 1615. Theoutput of the Bessel filter (V-RECT) is provided to a low pass filter(approximately 2 Hz), which includes resistors 1611 and 1612 (68.1Kohms), capacitor 1604 (0.22 μF), capacitor 1605 (47 μF), and op amp1616. The output of op amp 1616 gives an average of the input linevoltage (V-LINE).

V-LINE is provided to a typical precharge circuit 1625 which sets adelay before the electrolytic capacitors in the power supply precharge.An op amp 1626, and resistors 1629 (100K ohms) and 1630 (10K ohms),don't allow a capacitor 1627 (10 μF) to charge through a resistor 1628(100 ohms) until the line voltage reaches a threshold. After the linevoltage reaches the threshold capacitor 1627 charges to a level where itturns on a relay (not shown) through associated components including aresistor R63 (200K ohms), a resistor R51 (100K ohms), a resistor R108(619K ohms), an op amp U1, a diode D57, a NAND gate U2, and a resistorR89 (4.7K ohm). These components operate in a typical fashion. The relayenergizes and fires an SCR that precharges the electrolytic capacitors.

A multipliers/divider 1631 of receives the rectified line voltagesignal, and divides that by the average input (typically either 230 or460) so that a scaled rectified voltage is provided. Then the scaled, isrectified voltage is multiplied by an error signal from the bus toproduce a reference command. Specifically, V-RECT, the output of op amp1615 which corresponds to the rectified input voltage, is providedthrough a resistor 1632 (100K ohms) and an op amp 1633 as one channelinput to the multiplication. The other channel input to themultiplication is a BUS-ERROR signal provided through an op amp 1636A.

The output of op amp 1633 is provided through a log transistor 1635, andthe average line voltage (V-LINE) is provided through an op amp 1636 andto a log transistor 1637. The common junction between transistors 1635and 1637 is a subtraction, so the base of transistor 1637 is the resultof the subtraction. That difference is added through a transistor 1638to the bus error. The sum is provided to a transistor 1639, which takesthe anti-log of that value. Thus, a division and multiplication areperformed. The output is scaled by an op amp 1641 and associatedcircuitry including a diode 1642, a capacitor 1643 (0.001 μF), and aresistor 1645 (20K ohms).

A transistor 1646 limits the output current of op amp 1641, and iscontrolled by a resistor 1647 (20K ohms) and a diode 1648. The input toop amp 1650 is a scaled bus voltage, and sets the maximum outputcommand. The output command (VCOMM) is used to force the current shapeto match the input voltage shape.

The BUS-ERROR signal is provided by a typical error circuit whichincludes an op amp 1651 and associated circuitry resistors 1653 (20Kohms), 1654 (11K ohms), and 1655 (499K ohms), a diode 157 and acapacitor 1658 (0.047 μF). An 8 volt reference signal is compared to thedivided down (and scaled) 800 volt bus. An error signal is providedthrough a resistor 1659 (82.5K ohms) to op amp 1636A commanding anincrease or decrease in the current to increase or decrease the busvoltage. Also, the current command is adjusted by the shape of the inputsignal as provided through V-RECT to mimic the shape of input rectifiedsignal. Therefore, the current needed to result in a desired bus voltageis provided, but in such a shape that a power factor very close to oneis obtained.

The command signal is summed with a current feedback signal from a CT1by an op amp 1670 and provided to a boost drive circuit through logicgates (not shown) to turn on and off the IGBT in the preregulator. A CTis used to provide current feedback (rather than an LEM for example)because if a LEM fails it will call for unlimited current.

The boost drive signal is a digital signal of either zero (IGBT ON) orfifteen volts (IGBT OFF). The boost drive input is provided to the baseof a pair of transistors because the logic gates output do not provideenough current to drive the IGBT's. Thus the transistors providesufficient current. A transistor level shifts. The gates of a pair oftransistors are tied together by a capacitor (0.1 μF).

Another aspect of this invention is implemented with a half-bridge,transformer isolated, inverter that is SVT switched. The invertor uses aswitch circuit 1400, shown in FIG. 14, that includes a pair of switchesor IGBT's 1402 and 1403, and a pair of diodes 1404 and 1405. Diode 1404is an anti-parallel diode for switch 1402. Diode 1405 is ananti-parallel diode for switch 1403. The two switch/diode parallelcombinations are in series, but reversed, i.e. in opposing directions.This configuration provides a diode-type switch whose direction can bereversed.

An invertor using switch circuit 1400 is shown in FIG. 15, and includesa dc voltage source 1501, a pair of switches 1502 and 1504, with a pairof anti-parallel diodes 1503 and 1505, a pair of capacitors 1507 and1508 (1410 μF), a transformer 1509, a capacitor 1512 (0.099 μF), anoutput rectifier including diodes 1510 and 1511, and an output inductor1513.

Capacitor 1512 is switched across transformer 1509 by switches 1502 and1504. Switches 1402 and 1403 are used to soft switch switches 1502 and1504. Switches 1402 and 1403 do not need any special timing, and runwith the main clock at effectively 50% duty cycle. For example, switches1502 and 1402 turn on together, and switch 1502 delivers current totransformer 1509, while switch 1402 does nothing. When switch 1502 turnsoff, switch 1402 remains on, and current is directed through switch 1402and diode 1405 into capacitor 1512, thus giving an SVT (Slow VoltageTransition) turn off. Switch 1402 is turned off after the transition anddiode 1405 prevents the back flow of current from capacitor 1512. Thisoccurs in complimentary fashion with switches 1502 and 1402 and diode1405. Thus, this circuit provides full-wave transformer usage, PWMcontrol, complete capacitor balance control with no extra circuitry, andefficient use of switches with SVT.

Referring now to FIGS. 17-24, the various current paths followed duringa complete cycle are shown. The circuit in these Figures is analternative embodiment that includes splitting capacitor 1512 into twocapacitors, one connected to the upper bus, and one connected to thelower bus. This is done because the path through capacitors 1507 and1508 can reduce the effectiveness of the snubber substantially.

Initially, all of the switches are off in the snubber, and capacitors1512A and 1512B split the bus. The 800 volts bus is also split bycapacitors 1507 and 1508 (for half bridge operation). Capacitors 1507and 1508 should be large enough to keep the junction voltage betweenthem substantially constant during operation. Switches 1504 and 1402 areturned on together. Switch 1504 delivers power to transformer 1509 whileswitch 1402 is blocked by diode 1404. Thus, switch 1402 is in “standby”until switch 1504 turned off, as shown in FIG. 17.

Switch 1504 turns off, and the current through the transformer transfersto switch 1403, diode 1404 and capacitor 1512 (which form the snubberpath). The voltage across switch 1504 rises slowly giving a slow voltagetransition. This current path is shown in FIG. 18.

When the voltage across switch 1504 reaches the bus voltage, theremaining energy from transformer 1509 is spilled back into the busthrough diode 1503. This current path is shown in FIG. 19. When theremaining energy and transformer 1509 has been provided to the bus thesystem comes to rest with snubber capacitor 1512 fully charged tocompletely soft switch 1502 (FIG. 15).

After the system comes to rest switches 1502 and 1403 are turned ontogether. Switch 1502 delivers power to transformer 1509 while switch1403 is blocked by diode 1404. Thus, switch 1403 is in standby untilswitch 1502 is turned off (FIG. 20).

Switch 1502 is turned off and current from transformer 1509 transfers tocapacitor 1512 through the snubber path including diode 1404 and switch1403 rises slowly, giving a slow voltage transition. This current pathis shown in FIG. 21. The current continues in this path until thevoltage across switch 1502 has reached the bus, and remaining energy intransformer 1509 is spilled into the bus through diode 1505 (FIG. 22).Capacitor 1512 fully charged so that switch 1504 may be soft switched.The process then repeats.

One feature of the switched snubber used in FIGS. 15-22 is that the mainswitches (1504 and 1502) do not incur actual losses if the output poweris less than necessary to transition snubber capacitor 1512 from “railto rail”. Thus, it is not necessary to fully transition the snubber. Thereversible single direction switch prevents snubber interference on turnon, and thus provides snubbing proportional to load. This feature allowsvery heavy snubbing without restricting the load range of the inverter.

An alternative embodiment includes using a full bridge version of thesnubber.

FIG. 16 shows a control circuit for controlling the switching of theswitched snubber in FIGS. 14-22. Four gates drives 1402A, 1403A, 1502A,and 1504A are used to provide the gate signals for switches 1402, 1403,respectively. These gate drives are not shown in detail and areconventional gate drives such as those found in the Miller XMT 304®. Thegate drives are inverting in that a high input maintains the gates offand a low input maintains the gates on.

Gate drivers 1402A, 1403A, 1502A, and 1504A are controlled by a logiccircuit 2301. Logic circuit 2301 includes a plurality of NAND and ORgates in the preferred embodiment, however it's specific constructionmay be any of the designer's choosing. And enable signal is included asan input to logic circuit 2301, in one embodiment. They enable signal isused only during power down.

An error amplifying circuit 2303 is also shown in FIG. 16. Erroramplifying circuit 2303 may be a standard error circuit and is used, inthe preferred embodiment, with a CT feedback signal. The output of erroramplifier circuit 2303 is a PWM reference signal. The PWM referencesignal control is provided through an opto-isolator 2305 to electricallyisolate the remaining portion of the circuit from the error amp circuit.A pair of resistors 2306 (10K ohms) and 2307 (2K ohms) scale the PWMreference command for input into opto-isolator 2305. The output ofopto-isolator 2305 is scaled from a current to a voltage by a resistor2308 (10K ohms).

Generally, the control circuitry implements a modified PWM controlscheme. Above a minimum pulse width operation is a typical PWM scheme,and the pulse width is adjusted to increase or decrease current.However, for current less than that corresponding to the minimum pulsewidth the frequency of pulses is reduced (thus increasing the OFF time).The minimum pulse width is used because the gate drives have a limitedspeed.

The conventional pulse with modulation portion works with a ramp createdby an op amp 2310, resistors 2311 (10K ohms), 2312 (10K ohms), and 2313(200K ohms). The PWM reference command is received by op amp 2310through a diode 2314. The appropriate switch is turned on at the startof the ramp. The ramp is initiated by an op amp 2315 and resistors 2316(10K ohms), 2317 (611 ohms), 2318 (20K ohms), 2319 (200K ohms), 2321(6.11K ohms) and 2322 (2K ohms).

The main power switches (1502 and 1504) are maintained on for 95% of thetotal ramp time. The 95% threshold is set by an op amp 2325 and aresistor 2326 (10K ohms). The switches are turned off by changing stateson the set input of a flip flop 2327 (which is connected to op amp2325). The snubber switches (1402 and 1403) are switched off at 100% ofthe ramp.

A current source including transistors 2336 and 2331 and resistors 2333(332 ohms), 2334 (100 ohms) and 2335 9100 ohms). The current source setsthe slop of the ramp. When a capacitor 2337 (100 pF) discharges to athreshold set by a diode 2338 the ramp is restarted. The ramp willcontinue up at the slope set by the current source until the capacitorvoltage reaches the threshold set by op amp 2315 and its circuitry.

A flip flop 2328 is used to alternate between switches, and to receivean enable signal and a machine on/off signal.

Generally, the circuit operates as follows: the capacitor voltage isfully reset down to the minimum and then the ramp begins to ramp up andthe voltage on the capacitor is increasing from the current source. Asthe capacitor charges the output of the opto-isolator is provided to opamp 2310, which pulse width modulates the switches. When the capacitorvoltage rises above the reverence voltage set by the opto-isolator, opamp 2310 changes state, causing the switch to be turned off. Steeringflip flop 2328 determines which one, and only one, of the main powerswitches are on in a conventional manner. If the capacitor voltageincreases to the level set by op amp 2315 (95% of the peak), then themain power switch that is on is turned off.

The frequency adjust (for low current commands) operates as follows: theoutput of op amp 2315 (the ramp reset) is fed back through a NAND gate2341 through a resistor 2342 (100K ohms) and a buffering transistor2343. The machine on/off signal is also provided to transistor 2343. Theoutput of NAND gate 2341 also causes flip flop 2328 to change statethrough the clock input.

A voltage divider including resistors 2342 and 2345 (68.1K ohms) is tiedto a diode 2346. If diode 2346 pulls down the voltage at one end ofresistor 2345, then the voltage across a resistor 2347 (5.11K ohms) isalso pulled down. A current mirror including resistors 2349 and 2350(100 ohms) and transistors 2351 and 2352 provide the rest current forthe ramp. However, if the voltage through 2346 is low enough, then thevoltage input to transistor 2343 will be ground, and transistor 2343will not provide current to the current mirror to reset capacitor 2337,thus allowing the ramp to continue upward.

The various aspects of this invention, while described in the context ofa welding power supply has applications in many different areas.Generally, in applications where low loss switching is desirable using aboost convertor this arrangement may be used.

Numerous modifications may be made to the present invention which stillfall within the intended scope hereof. Thus, it should be apparent thatthere has been provided in accordance with the present invention amethod and apparatus for providing power with a high power factor andlow switching losses that fully satisfies the objectives and advantagesset forth above. Although the invention has been described inconjunction with specific embodiments thereof, it is evident that manyalternatives, modifications and variations will be apparent to thoseskilled in the art. Accordingly, it is intended to embrace all suchalternatives, modifications and variations that fall within the spiritand broad scope of the appended claims.

What is claimed is:
 1. A welding power supply comprising: an inputrectifier configured to receive an input line voltage and provide arectified voltage on an output; a pre-regulator connected to receive asan input the output of the rectifier and provide a dc bus as an output;and an invertor, connected to receive the output of the pre-regulatorand provide a welding output; wherein the inverter includes a snubbercircuit having a first switch in anti-parallel with a first diode, and asecond switch in anti-parallel with a second diode, and wherein thecombination of the first switch and first diode are connected in serieswith the combination of the second switch and the second diode, andwherein the first and second switches are connected in opposingdirections.
 2. A welding power supply comprising: a first current paththrough a transformer in a first direction, the first current pathincluding at least a first switch with an anti-parallel first diode; asecond current path through the transformer in a second direction, thesecond current path including at least a second switch with ananti-parallel second diode; a snubber, including a current path having athird switch with an anti-parallel third diode, a fourth switch with ananti-parallel fourth diode, wherein the third switch and ananti-parallel diode are in series with, and oppositely directed from,the fourth switch and anti-parallel diode, and at least one snubbercapacitor.
 3. The power supply of claim 2 wherein the first and secondswitches are in a half-bridge configuration.
 4. The power supply ofclaim 2 wherein the at least one snubber capacitor includes a firstsnubber capacitor connected with a first bus line and a second snubbercapacitor connected with a second bus line.
 5. The power supply of 2wherein the at least one snubber capacitor is in series with the thirdand fourth switches and anti-parallel diodes.
 6. A method of providingwelding power from a welding power supply including a first power switcha second power switch, a first snubber switch, a second snubber switch,a first dc bus, a second dc bus, a first power capacitor, a second powercapacitor, a first snubber diode, a second snubber diode, a firstsnubber capacitor, a second snubber capacitor, a first anti-parallelpower diode, a second anti-parallel power diode, and a transformer,comprising: turning on the first power switch and the first snubberswitch, and allowing current to flow through the first power switch, thefirst dc bus, the first power capacitor, and in a first directionthrough the transformer; turning the first power switch off and allowingcurrent to flow through the first snubber switch, the second snubberdiode, the first snubber capacitor, and through the transformer in thefirst direction, while the first power switch is turning off, to providea slow voltage transition off; allowing current to flow through thesecond anti-parallel power diode, the second DC bus, the second powercapacitor, and through the transformer in the first direction, while thefirst power switch is continuing to turn off, to continue providing aslow voltage transition off; turning off the first snubber switch;turning on the second power switch and the second snubber switch afterthe first power switch is off, and allowing current to flow through thesecond power switch, the transformer in a second direction, the secondpower capacitor, and the second dc bus; turning the second power switchoff and allowing current to flow through the second snubber switch, thefirst snubber diode, the transformer in the second direction, and thesecond snubber capacitor, while the second power switch is turning off,to provide a slow voltage transition off; allowing current to flowthrough the first anti-parallel power diode, the transformer in thesecond direction, and the first power capacitor, while the second powerswitch is turning off, to provide a slow voltage transition off; turningoff the second snubber switch; and repeating these steps.
 7. The methodof claim 6, wherein turning on the first and second power switchesincludes soft switching on the first and second power switches.
 8. Amethod of providing welding power from a welding power supply includinga first power switch a second power switch, a first snubber switch, asecond snubber switch, a dc bus, a power capacitor, a first snubberdiode, a second snubber diode, as snubber capacitor, a firstanti-parallel power diode, a second anti-parallel power diode, and atransformer, comprising: turning on the first power switch and the firstsnubber switch, and allowing current to flow through the first powerswitch, the dc bus, the power capacitor, and in a first directionthrough the transformer; turning the first power switch off and allowingcurrent to flow through the first snubber switch, the second snubberdiode, the snubber capacitor, and through the transformer in the firstdirection, while the first power switch is turning off, to provide aslow voltage transition off; allowing current to flow through the secondanti-parallel power diode, the DC bus, the power capacitor, and throughthe transformer in the first direction, while the first power switch iscontinuing to turn off, to continue providing a slow voltage transitionoff; turning off the first snubber switch; turning on the second powerswitch and the second snubber switch after the first power switch isoff, and allowing current to flow through the second power switch, thetransformer in a second direction, the second power capacitor, and thedc bus; turning the second power switch off and allowing current to flowthrough the second snubber switch, the first snubber diode, thetransformer in the second direction, and the snubber capacitor, whilethe second power switch is turning off, to provide a slow voltagetransition off; allowing current to flow through the first anti-parallelpower diode, the transformer in the second direction, and the powercapacitor, while the second power switch is turning off, to provide aslow voltage transition off; turning off the second snubber switch; andrepeating these steps.
 9. The method of claim 8, wherein turning on thefirst and second power switches includes soft switching on the first andsecond power switches.
 10. The method of claim 9, wherein the powercapacitor is comprised of first and second capacitors, and the dc bus isa split bus.
 11. The method of claim 9, wherein the snubber capacitor iscomprised of first and second capacitors, and the dc bus is a split bus.